Efficient generation of radio frequency currents

ABSTRACT

In the present invention, pre-distortion of drive signal and generation of bias signal to a power amplifier are both controlled dependent on an instantaneous size of the input signal, for producing a predetermined gain characteristics. Preferably, the bias signal is kept low in amplitude ranges having a high probability to occur, thus giving a high efficiency, and is allowed to increase towards higher amplitudes, preferably all the way to the maximum amplitude. The pre-distorted drive signal is preferably higher than the input signal in the high-efficiency ranges. Preferably, the drive signal is predominantly composed of low-order components. In cases where signal paths of bias signal and drive signal differs significantly, inverse filtering is applied to ensure the simultaneousness at the input of the amplitude element.

This application is the U.S. national phase of international applicationPCT/SE03/00458 filed in English on 19 Mar. 2003, which designated theU.S. PCT/SE03/01068, and claims priority to SE Application No. 0201908-1filed 19 Jun. 2002. The entire contents of these applications areincorporated herein by reference.

TECHNICAL FIELD

The present invention relates generally to the field of power amplifiersand in particular to efficiency enhancement for power amplifiers.

BACKGROUND

In many wireless communication systems, a power amplifier (PA) used inthe transmitter is required to be very linear. High linearity isrequired to prevent leakage of interfering signal energy between theintended channels. In addition, it has to allow for simultaneousamplification of many radio channels or frequencies spread across afairly wide bandwidth. In order to reduce power consumption and need forcooling, the amplifier also has to have a high efficiency.

It is difficult to provide linear RF currents with high efficiency forpractical high-power RF transistors. Practical high-power RF transistorshave a transconductance (output current per input voltage) that is notconstant, but changes with the input node voltage. A conventional way ofgetting a linear RF current response with high DC-to-RF conversionefficiency is to bias the device in a so-called class B operation (forclass definitions, see e.g. [1]), where an output current is providedduring half of each RF cycle. The biasing is performed by providing a DCvoltage offset at the input node, such as the gate of a field-effecttransistor (FET) device or the base of a bipolar transistor. If thetransconductance of a class B device ideally is constant for positiveinput signal voltages and zero for negative input signal voltages, theRF output current is linear and the maximum efficiency is 78.5% asdiscussed e.g. in [2]. Since the DC current consumption is proportionalto the RF current, the power efficiency of a class B amplifier isproportional to the output amplitude, provided that the drain orcollector supply voltage is kept constant.

In real RF transistors, the transconductance may differ significantlyfrom constant. It can e.g. be a substantially linear function of theinput node voltage, or a mixture of a mostly linear part and a mostlyconstant part. This behavior is normally referred to as quasi-linear. Inthe case of a purely linear transconductance, the output signalamplitude is proportional to the input signal squared for a class Bbiased amplifier. Such response is sometimes referred to as a parabolictransfer function.

Since RF power transistors typically are relatively expensive, there isalso an incentive to get the most possible output power from them. Animportant factor in getting the most output power from a transistor isto have the right load impedance at the fundamental frequency, i.e. thedesired RF. Practical transistors have limits to both the maximum outputcurrent and the maximum output node voltage. To obtain the maximumoutput power from the device, it is therefore important that theselimits are reached simultaneously.

Besides the above-described class B biasing, there are also otherbiasing possibilities. An amplifier biased in class A is always givingan output current. An amplifier in class AB is biased between class Aand class B. A class A amplifier gives the highest output RF power for agiven limitation in the maximum peak current, when most devices withlinear or quasi-linear transconductance are used. For a device with aconstant transconductance, biasing in class A and class B gives the sameout-put power, and class AB slightly more. With a tuned load, i.e. allvoltage harmonics at the output node perfectly short-circuited or“terminated” by a reactance network, and a device with a lineartransconductance, class B operation gives 0.7 dB less output power thanclass A, see e.g. [2]. By terminating the voltage harmonics, inparticular the second harmonic, at the output node makes it possible touse a larger voltage swing of the fundamental frequency. However, theharmonic termination is seldom perfect in practice. Since the harmoniccontent of the class B current waveform generally is greater than thatof class A, the difference in output power can be even larger than 0.7dB.

Biasing in class C means that the RF current is on during less than halfof each RF cycle. This biasing gives generally the highest efficiency,but gives highly nonlinear response and considerably less peak outputthan either class A or B. In order to maximize the peak output, class Abiasing is required, but in order to maximize the efficiency, class C isto prefer. In addition to this linearity considerations have to betaken.

A simple way of getting a linear response from a device with lineartransconductance is to bias it in class A. This gives, however a verylow average efficiency if the peak-to-average power ratio is high, dueto the high quiescent DC supply current. A quasi-linear device can bebiased in class AB, and have a substantially linear response for lessquiescent current. The average efficiency is therefore better than fordevices with linear transconductance biased in class A, but worse thandevices with constant transconductance biased in class B. In order toachieve a better linearity, the bias level can be controlled adaptively.Such bias control is generally much slower than the signal variations.Solutions using adaptive bias levels are discussed e.g. in [3-7].

In many cases, bias level adjustments, i.e. choosing the best staticbias, are not enough to obtain sufficient linearity. Furtherlinearization methods must typically be employed, see e.g. [5-7]. Oneoften used method for providing high efficiency and wide bandwidth ispre-distortion, used e.g. in [5, 7]. The input signal to the transistoris nonlinearly pre-compensated to counteract the distorting behavior ofthe transistor itself. In some cases, the linearizing performance of thepre-distortion method is even sufficient to allow for the device to bebiased closer to class B, which normally implies a higher degree ofnon-linearity. However, the quiescent current is reduced and the averageefficiency is increased.

Another way of getting a more linear response from a practical device isto use a dynamic bias. This means that the bias level is variedsubstantially at the same speed as the amplitude of the amplifiedsignal. Examples of such systems are found in [8-13]. In [8], suchtechnique has been used e.g. for improving the power-added efficiency ofclass A amplifiers with low gain. A constant gain device is used, whichcan be kept in class A with a bias level proportional to the signalamplitude.

Two different dynamic biasing schemes optimizing either efficiency orintermodulation distortion are compared in [9]. In [10], it is proposedto take care of extra non-linearity introduced by dynamic biasing byapplying pre-distortion. Using dynamic biasing to provide constant gain(and therefore linear amplification) with higher efficiency than class Aoperation is described in [11], and with the addition of feedback forcontrolling the bias level, in [12]. In [11], the device is biased inclass AB at low amplitudes, and ends up in class B at high amplitudes,and in [12] the device is biased in class AB, and the bias variationsare only used for small corrections by feedback to the momentary gain.Phase distortion is handled by a separate feedback loop. Constant gainis also the objective in [13], in which a dynamic bias scheme is addedonto a Doherty high-efficiency amplifier.

[14] discloses a bias circuitry configured to adjust bias current to oneor more power amplifier stages based upon the level of the RF signal tobe amplified. The bias circuitry eliminates excessive quiescent biascurrents that prior biasing technique required to ensure linearoperation by automatically increasing bias currents only as needed basedon the effective magnitude of the RF signal to be amplified.

In [15] a power transistor circuitry is disclosed, which for increasingthe gain of an amplified RF signal has a power transistor, a voltagebias circuit having a peak detector, and a voltage source. A summer sumsthe outputs of the peak detector and the voltage source to increase thebias and provide an increased gain upon detection of an RF signal peak.

SUMMARY

A general problem with prior-art power amplifier arrangements is theinability to obtain a good linearity, high output power and highefficiency simultaneously. A further problem is that highly controllablearrangements typically are based on solutions employing large bandwidth,which puts high demands on D/A converters and thereby increases thecosts for the devices.

An object of the present invention is to provide devices and methodsenabling a better controlled compromise between linearity, high outputpower, high efficiency and narrow bandwidth. A further object of thepresent invention is to enhance the efficiency for signal amplitudeshaving a high probability to appear. Another object of the presentinvention is to allow for tailoring output signal characteristicsdepending on the amplifier application.

The above objects are achieved by devices and methods according to theenclosed claims. In general words, generation of bias signal to a poweramplifier is controlled dependent on the instantaneous size of the inputsignal for producing a predetermined output characteristics. The biassignal is selected for efficiency reasons and increases thenonlinearilty of the actual amplifying step. Such nonlinearity ispreferably compensated by adapting the drive signal provided to theamplification step. Such modifying comprises preferably pre-distortionof the drive signal or feedback arrangements. Preferably, the biassignal is kept low in amplitude ranges having a high probability tooccur, thus giving a high efficiency. Amplification according to class Bor class C is used in this region. In order to have access to as highoutput power as possible, the bias signal is allowed to be higher closeto the maximum amplitude, i.e. higher than for a class B amplification.The drive signal amplitude curve is preferably adapted to provide alinear output signal. The adapted drive signal has preferably a higherderivative than the input signal in the low high-efficiency ranges.Preferably, the drive signal is predominantly composed of low-ordercomponents. In cases where signal paths of bias signal and drive signaldiffers significantly concerning delay and frequency response, inversefiltering and/or delay compensation is applied to ensuresimultaneousness at the input of the amplitude element.

One of the main advantages with devices and methods according to thepresent invention is that the increased controlling possibilities ofboth drive and bias signals gives an additional freedom in controllingthe power amplifier operation. This additional freedom can be used indifferent applications to enhance certain requested properties, e.g.linearity, efficiency, low bandwidth etc.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention, together with further objects and advantages thereof, maybest be understood by making reference to the following descriptiontaken together with the accompanying drawings, in which:

FIG. 1 is a schematic drawing of a transmitter having a radio frequencypower amplifier;

FIG. 2 is a schematic drawing of a wireless communication system;

FIG. 3 is a diagram illustrating drive signal and bias signal providedto an amplifier element as a function of input signal amplitude,according to an embodiment of the present invention;

FIG. 4 a is a diagram illustrating drive signal and bias signal providedto an amplifier element as a function of input signal amplitude,according to a prior-art pre-distorting device;

FIG. 4 a is a diagram illustrating drive signal and bias signal providedto an amplifier element as a function of input signal amplitude,according to a prior-art device using dynamic bias;

FIG. 5 is a diagram illustrating the efficiency of the embodiments ofFIG. 3, FIG. 4 a and FIG. 4 b.

FIG. 6 is a diagram illustrating the probability amplitude distributionfor a typical multi-carrier signal;

FIG. 7 a is a diagram illustrating the frequency distribution of a drivesignal according to the embodiment of FIG. 4 a;

FIG. 7 b is a diagram illustrating the frequency distribution of a biassignal according to the embodiment of FIG. 4 b;

FIG. 7 c is a diagram illustrating the frequency distribution of a drivesignal according to the embodiment of FIG. 4 c;

FIG. 7 d is a diagram illustrating the frequency distribution of a biassignal according to an embodiment of the present invention;

FIG. 7 e is a diagram illustrating the frequency distribution of a drivesignal according to an embodiment of the present invention;

FIG. 8 is a block diagram of an embodiment of a power amplifieraccording to the present invention;

FIG. 9 is a block diagram of a preferred embodiment according to thepresent invention for providing control of drive and bias signals;

FIG. 10 is a block diagram of a Doherty amplifier arrangement, in whichan amplifier according to the present invention is useful;

FIG. 11 is a diagram illustrating drive signal and bias signal providedto an amplifier element as a function of input signal amplitude,according to another embodiment of the present invention; and

FIG. 12 is a flow diagram illustrating the main steps in an embodimentof a method according to the present invention.

DETAILED DESCRIPTION

In a radio frequency application for power amplifiers, as illustratede.g. in FIG. 1, a power amplifier 2 is arranged in a radio transmitter10 for amplification of several narrow-band channels simultaneously. Theradio transmitter 10 comprises in this embodiment of a general inputcombination unit 6, receiving input signals from a number of sources,each one representing a channel, carrier etc. The input signals arecombined in the general input combination unit 6 into a multi-channelsignal. The multi-channel signal is provided to the power amplifier 2for simultaneous amplification. The amplified signal is finally providedto a transmission element 4. Alternatively, base-band signals areprovided as input signals and an up-conversion to the radio frequency isadditionally provided. The details of the operation are omitted, sincethey are performed according to conventional techniques and do notconcern the essential parts of the present invention.

In order to preserve the phases and amplitudes of all signal componentsin the amplification process and to prevent leakage of interferencesignal energy into frequency bands outside the channels, a high degreeof linearity is required in such an amplifier.

Power amplifiers can be found in many fields of technology. Theapplications may range from consumer electronics, radar technology toradio communication. The transmitter illustrated in FIG. 1 is assumed tobe a part of a radio communication system. It should though beunderstood that the invention is not limited thereto, and that otherapplications are feasible as well. However, references to applicationswithin the field of radio communication will be made in the presentdisclosure for serving as examples.

An embodiment of a radio communication system 1 is illustrated in FIG.2. The system comprises a number of cells 5, covering a certain systemarea. Each cell 5 has an associated base station 9, which provides forradio communication within cell boundaries, represented by circles inFIG. 2. In this particular embodiment, the radio communication isprovided by omni-directional transmitters. A mobile unit 8 being presentin one of the cell communicates with the base station 9 of that cell.For reducing complexity of the illustration, only one base station 9 andone mobile unit 8 have been illustrated in the figure. The base station9 and/or the mobile unit 8 may be equipped with a transmitter equipment,e.g. according to FIG. 1, and the power amplifier is preferably designedaccording to the present invention.

In FIG. 8, an embodiment of a power amplifier 2 is illustrated. Thepower amplifier 2 comprises a input signal terminal 53, on which a inputsignal 35 is received. This signal is intended to be amplified andprovided as a radio frequency output signal 55. The input signal 35 isin this embodiment a digital signal representing a baseband version ofthe signal to be transmitted. The input signal terminal 53 is connectedto a pre-distortion unit 20, which pre-distorts the input signal 35 intoa drive signal 26. This pre-distortion will be described more in detailfurther below. The pre-distortion unit 20 is in this embodiment adigital pre-distortion unit 20.

The input signal terminal 19 is also connected to an input detector 40,which is arranged to determine an instantaneous (envelope) size measureof the input signal 35. The size measure is preferably the power of theinput signal 35 or a quantity derivable therefrom, e.g. amplitude. Theinput detector 40 has an output connected to the pre-distortion unit 20,whereby the pre-distortion unit 20 can apply a pre-distortion dependenton an signal 47 representing the instantaneous amplitude of the inputsignal 35.

An output of the input detector 40 is also connected to a bias signalgenerator 30, which generates a bias signal 36 to be used for providingthe actual amplification operation. The bias signal generator 30 canthus provide a bias signal 36 dependent on an signal 47 representing theinstantaneous amplitude of the input signal 35. In the presentembodiment, the bias signal generator 30 is digital.

In the present embodiment, the input detector 40 is indicated as aseparate unit. However, the input detector 40 can also be implemented asa part of the pre-distortion unit 20 and/or the bias signal generator30, respectively.

The drive signal 26 is in the present embodiment processed in an inversefilter device 21, described somewhat more in detail further below. Theinverse filter device 21 is in this embodiment digital. The filtered(and/or delay compensated) signal is brought further to adigital-to-analogue converter (D/A) 22 for the provision of an analoguesignal. The signal is then in this embodiment modulated to aradio-frequency a RF chain 23.

The bias signal 36 is according to the present embodiment similarlytreated in an inverse filter device 31. The filtered (and/or delaycompensated) signal is then brought further to a digital-to-analogueconverter (D/A) 32 for the provision of an analogue bias signal, whichsubsequently is amplified in a bias amplifier 34.

The drive signal and the bias signal are matched in a matching unit 25and provided to an amplifier element 50, performing the actualamplification according to the selected drive signal and bias signal. Aradio frequency output signal 55 is thus provided.

The drive signal passes through a number of steps in a drive signal path51, between the inverse filter device 21 and the matching unit 25. Thesignal processing, digital and/or analogue, along the drive signal path51 may influence the delay and frequency response of the drive signal.Similarly, the bias signal also passes through a number of steps in abias signal path 52, between the inverse filter device 31 and thematching unit 25. The signal processing, digital and/or analogue, alongthe bias signal path 52 may influence the frequency responses and delayof the bias signal. If response and delay differences along thedifferent signal paths 51, 52 differs, which is the case in a typicalamplifier, the drive signal arriving at the amplifier element 50 doesnot correspond to the bias signal arriving at the same time, unlessinverse filtering and/or delay compensation is performed. Suchcompensation or means for causing simultaneousness comprises in thepresent embodiment the two inverse filter devices 21 and 31,respectively. The parts mostly responsible for frequency dependenciesare the radio frequency chain, which typically includes several filters,the radio frequency match and the bias network and the bias amplifier.The inverse filter device 21 causes distortion and/or delay compensationof the drive signal, which is inverse to the distortions and delayscaused during the signal path 51, thereby the name. It means that whenthe signal has passed both the inverse filter 21 and the signal path 51,substantially all distortions have cancelled and delays are equal.Similarly, the inverse filter device 31 causes a distorted frequencyresponse and delay compensation of the bias signal that is inverse tothe distortions and delays caused during the signal path 52. Itanalogously means that when the bias signal has passed both the inversefilter device 31 and the signal path 52, all distortions have cancelledeach other. The safest way to accomplish that the drive and bias signalsare simultaneous at the input node is to inverse-filter both parts. Thistakes away the frequency dependencies for the nonlinearly processedsignals within the bandwidth of the inverse filters.

In particular in high-linearity wideband applications, it is importantthat the bias and pre-distorted drive signal coincide at the input nodeof the amplifier element.

The power amplifier 2 of the illustrated embodiment also comprises afeed-back arrangement 48. The radio frequency output signal 55 is sensedby a feedback sensor 41. The feedback sensor 41 also preferably performsa radio-frequency down-conversion. The sensor signal is processed in ananalogue-to-digital (A/D) converter providing a signal 46 representingthe radio frequency output signal 55. This signal 46 and the inputsignal are provided to an adaptation unit 44. The adaptation unit 44modifies the operation of the bias signal generator 30 and thepre-distortion unit 20 by supplying coefficient modifications ofcoefficients stored in the look-up tables 53, 54. The bias signalgenerator 30 and the pre-distortion unit 20 are arranged to adapt theiractions accordingly.

From FIG. 8 it is easily seen that there are means for detecting theinstantaneous amplitude of the input signal and to use this informationfor controlling the bias signal as well as the pre-distortion of thedrive signal in order to achieve an output signal of requestedproperties.

The output of an amplifier element, e.g. a field-effect transistor or abipolar transistor, is basically dependent on the drive signals, a biassignal and the inherent amplification properties of the amplifierelement. According to the present invention, both the drive signal andthe bias signal are controlled based on the amplitude of the inputsignal. In FIG. 3, drive signal 26 and bias signal 36 used in anoperation of a power amplifier according to an embodiment of the presentinvention based on a linear-transconductance amplifier element isillustrated. The amplitude of the drive signal 26 and bias signal 36 areplotted as functions of the amplitude of the input signal. An outputsignal 55 of the amplifier element is also illustrated. In thisparticular embodiment, one object is to keep the output signal 55 linearwith respect to the input signal. A linear amplitude range 13 covers inthis embodiment thus the entire amplitude range. As an additionalobject, the total power consumption of the arrangement is kept low andalso the bandwidth of the drive signal.

A power amplifier can be biased in many different ways. Operating apower amplifier as a so-called class A amplifier means that theamplifier is constantly conducting and delivers a instantaneous currentthat is higher than zero. To achieve this, the power amplifier is biasedto a DC voltage corresponding to the peak amplitude of the radiofrequency input voltage. This is the only way to get maximumradio-frequency current, and hence maximum output power for lineartransconductance and most cases of quasi-linear transconductancedevices. Well-adjusted class A biasing can give good linearity, butsometimes not enough for demanding applications. In such cases, furtherlinearization is needed. However, the quiescent current in class A islarge, which means that the efficiency becomes very poor. Adaptivequasi-static biasing, see e.g. [3-7] does not improve efficiency butprovides the adjustment to the class A amplifiers for improvinglinearity.

For keeping a good efficiency, but still having access to the maximumpower, the bias signal according to the present invention is preferablycontrolled to give a class A operation for the maximum amplitude.However, for lower amplitudes, the biasing level is reduced. In otherwords, there exists a first amplitude, such that the amplitude-averagedvalue of the bias signal for an amplitude interval over this firstamplitude is higher than the amplitude-averaged bias signal as takenover the entire amplitude range.

The average efficiency is not much affected by doing this, since thereis a low probability for the highest amplitudes to occur. The amplitudeprobability density of a mix of sufficiently many independent radiofrequency channels tends to be similar to a Rayleigh distribution,having a large peak-to-average power ratio. In FIG. 6, an example ofsuch a amplitude probability density distribution 15 is illustrated.Here it can be seen that a very small portion of the amplitudedistribution appears at or close to the maximum amplitude. A decreasedefficiency for the highest amplitudes will thus not affect the totalefficiency in any substantial manner.

When discussing bias levels in the present disclosure, these levels aregenerally given relative to a turn-on voltage of the power amplifier.

A class B operation corresponds to a biasing, which delivers half-waverectified current pulses, conducting during half a radio frequencycycle.

Class AB denotes the bias region between class A and class B. A furtherreduction of the biasing level, i.e. the introduction of negativebiasing potentials reduces the “on”-period further and so the totalefficiency increases. Such negative biasing is referred to as class C.

Returning to FIG. 3, the amplitude of the drive signal here consists ofa mixture of the first and third powers of the input amplitude, whichmeans that this signal can be contained within three times the bandwidthof the input signal. This will be discussed further below. The biassignal to the amplifier is dynamically changed according to the curveshown, for achieving a linear output signal with the employed drivesignal. Due to the substantial amount of class C operation, the averageefficiency is good.

Both the pre-distortion of the drive signal and the bias level can becalculated from the squared amplitude (i.e. normalized instantaneouspower) of the input signal, or by the envelope of the input signal, i.e.its amplitude.

For amplitudes differing considerably from maximum amplitude, it istypically unsuitable to use class A amplification, due to the loss inefficiency. From FIG. 3 it is easily seen that when reducing theamplitude from maximum amplitude, the bias signal decreases in thisembodiment through class AB operation, passes class B operation and inthe intermediate amplitude range, a class C operation is used. Thus, inan amplitude range 11, the bias signal increases with increasing inputsignal amplitude. Preferably, this increase continues all the way to themaximum amplitude in order to provide class A operation only atamplitudes close to the maximum.

A dotted line 16 indicates the mean bias level averaged over the entireamplitude range. In FIG. 3 it can be noted that the bias signal 36 iskept below this amplitude-averaged bias level in an amplitude range 12.The amplitude range 12 thus represents a range, where the efficiency ofthe amplifier is relatively high. By comparing FIG. 3 and FIG. 6, onerealizes that class C operation, i.e. within broken lines, as well asoperation within the range 12 (dotted lines) falls in an amplitude rangehaving a large portion of the total amplitude distribution. In otherwords, a large portion of all signals to be amplified fall within therange 12. This will increase the overall efficiency of the poweramplifier.

A constant bias amplifier is generally nonlinear. Prior art dynamic biassystems typically aims to remove or at least reduce this nonlinearity.When considering the actual amplification step in the present invention,i.e. comparing the drive signal and the output signal, the biasingaccording to the above ideas gives rise to an increased nonlinearity.Instead, this biasing concepts open up for a more efficient use of theamplifier. In applications where linearity is of importance, theincreased nonlinearity is instead compensated by adapting the drivesignal before the actual amplification.

Such drive signal adaptation is preferably performed by pre-distortion.However, other approaches to modify the drive signal, such as feedbackarrangements can also be used.

In FIG. 4 a, drive signal, output signal and bias signal for a class Bamplifier with pre-distortion according to prior art are illustrated asfunctions of amplitude. Similarly, in FIG. 4 b, drive signal, outputsignal and bias signal for an amplifier with dynamic bias according toprior art, e.g. [11], are illustrated as functions of amplitude. It canhere be noted that the dynamic bias signal presents a monotonicallydecreasing behavior, in sharp contrast to the increasing behavior inrange 11 (FIG. 3) discussed above. These two systems are in thefollowing used as comparison to the devices according to the presentinvention. The peak amplitude for the systems of FIG. 3, FIG. 4 a andFIG. 4 b are equal.

A solution according to the state of the art to the problem of biasingin or near class B when using practical devices is typically to usedynamic biasing to get a constant gain over the input range, as seen ine.g. [8, 11, 12, 13]. Three major problems are associated with this.First, for systems with high linearity requirements, constant gain isnot enough to ensure linearity. The amplifier also has an amplitude tophase distortion, which is not addressed by the dynamic biasing and thusneeds further linearization, i.e. pre-distortion such as in [10] orfeedback such as in [12].

The second problem is that constant-gain dynamic biasing excludes thepossibility of obtaining maximum power while maintaining a highefficiency. This is because the maximum output power is associated withoperation in pure class A, which also has the highest gain. Theamplifier must then, for operation with constant gain and maximum peakoutput power, be biased essentially always in class A. This degradesefficiency drastically.

The third problem is that efficiency is lower than with class B biasing,since the device is biased in class AB at low and medium levels [11, 12,13]. The difficulty of simultaneously obtaining high linearity and highefficiency is further illustrated in [9].

The conclusion is that the state-of-the-art solutions using dynamicbiasing have deficiencies in several ways, mostly associated withobtaining high efficiency together with high output power and goodlinearity. However, these deficiencies are easily overcome by devicesand methods according to the present invention.

In FIG. 5, the efficiency of a class B amplifier with constanttransconductance is plotted in a diagram as a function of inputamplitude as a dotted line. The efficiency of the system of FIG. 4 a isplotted in the diagram as a function of input amplitude as a broken linewith short sections. The efficiency of the system of FIG. 4 b is plottedin the diagram as a function of input amplitude as a broken line withlong sections. Finally, the efficiency of the system of FIG. 3 isplotted in the diagram as a function of input amplitude as a full line.From this, it is easily seen that both the dynamic bias system and thepure class B amplifier are less efficient than the pre-distortedamplifier and the present invention in those regions having the highestamplitude distribution probability. The average efficiencies, for asignal having a 10-dB peak-to-average Rayleigh amplitude distributionare 30% for class B with pre-distortion, 24% for dynamic biasing and 30%for the embodiment according to FIG. 3.

A large disadvantage with class B amplifier with pre-distortion, besidesthe problem of not obtaining maximum power, is that a large bandwidthhas to be used. A major cause of the large bandwidth expansion for theradio frequency drive signal in the pre-distorted class B amplifier isthe very low gain at low levels for linear transconductance devices orquasi-linear transconductance devices, which need to be counteracted bya substantially root function amplitude distortion, as further explainedhere below. The constant gain dynamic biasing solves this problem, butwith a severe penalty in efficiency, as seen above.

The trade-off involved in the choice of bias level (static orquasi-static) and predistortion is mainly the bandwidth expansion. Thecloser the circuit is biased towards class B, the more pronounced is thenon-linear response. For linear transconductance devices, this responsedevelops towards the square of the input signal amplitude. Thepre-distortion that counteracts this development therefore developstowards a root function in amplitude. Such a function has a very largebandwidth, i.e. a large bandwidth expansion, compared with the originalsignal. For a certain level of linearity in the output, the bandwidth ofthe predistortion circuitry must thus be made much greater for a devicebiased in class B than for devices biased more into the class AB region.Since high-precision pre-distortion is usually performed with a digitalprocessing, a wider pre-distortion bandwidth means that more widebanddigital-to-analogue (D/A) converters must be used. These converters aregenerally more expensive than more narrowband ones, and in some casesthe required combination of precision and bandwidth does not even exist,making it impossible to increase efficiency by biasing more towardsclass B with maintained linearity.

The solution to this, as offered by the present invention, e.g. FIG. 3,is to abandon constant gain biasing in favor of something that allowsfor greater efficiency. This can be accomplished by the earlierdemonstrated class C biasing in the region where the amplitude of amulti-carrier or multi-user signal spends most of the time, i.e. in themiddle region. This is combined with a boost in drive amplitude in thisregion, to counteract the lower gain coming from biasing in class C. Thenon-linear drive boost can be forced to occupy a much narrower bandwidththan the pre-distortion bandwidth of the class B amplifier, by mainlyallowing only an odd-powers, low-order modification.

The modified, substantially low-order, amplitude function of the drivesignal in FIG. 3 can not generally compensate for the low gain of classB, or even lower gain of class C biasing, at low amplitude levels. Thismeans that the amplifier must be taken out of this high-efficiency modein this amplitude region, which can be accomplished by biasing in classA or AB. The average efficiency loss for this is not very high, however,since the amplitude probability density generally is low also for lowamplitudes.

By comparing FIG. 3 and FIG. 4 a, it can be noticed that in FIG. 3, thedrive signal has a pronounced maximum at intermediate amplitudes, whilethe gradient at low amplitudes is kept relatively low. Such a shape ofthe pre-distortion is relatively easy to achieve by using low-ordercomponents. The sharp initial increase of the pre-distorted drive signalto the class B of FIG. 4 a requires a fairly high degree of high-ordercomponents.

In a comparison between the systems discussed above, it can be notedthat the system according to the present invention has at least 0.7 dBmore peak output power, which means that the optimum load resistance forsuch a system, peaking in class A, is 85% of that of the two othersystems, peaking in class B. This is because the peak voltage amplitudesare equal for the three systems, if they all have perfect termination ofthe harmonics and equal supply voltages. The harmonics of the currentwave form are considerably lower in the peak amplitude range for thesystem according to the present invention. The demands for terminationwill therefore also be reduced.

The bandwidth requirements are illustrated in FIGS. 7 a-e. The bandwidthexpansions of the processed drive signal for the three cases of FIG. 3,FIG. 4 a and FIG. 4 b are shown as well as the bandwidth of the biassignal. FIG. 7 a illustrates the pre-distorted class B amplifier, andthe root shape of the drive signal results in a broad bandwidthexpansion. The bias signal for this case is not shown, since it consistsof only DC or slowly adjusted DC, which needs only a very narrowband D/Aconverter. FIG. 7 b illustrates the bias signal of the the constant gaindynamic bias case. FIG. 7 c illustrates the corresponding drive signal,centered around a central radio frequency. The bias signal is dominatedby DC and a second order component in the amplitude. The bias signaltherefore has the same bandwidth as the input signal. However, the drivesignal bandwidth is significantly narrower. Finally, in FIGS. 7 d and 7e, bias signal and drive signal, respectively, of an embodiment of thepresent invention corresponding to FIG. 3 is illustrated. Here it isseen that the bias signal has significant components above order four,which means that it has at least twice the input signal bandwidth.However, the drive signal consists of predominantly first and thirdorder components, which means that the bandwidth expansion is reduced toabout three times the input signal bandwidth. The requirements on theD/A converters are thus kept at a relatively moderate level.

The above discussed bandwidth diagrams are only examples of possiblecases. Cases with better or worse bandwidth distribution than thediscussed ones may be provided.

The radio frequency and bias signal spectra show that this example ofthe new system according to the present invention in practice does notrequire more available D/A converter bandwidth for producing the radiofrequency signal than the constant gain dynamic bias scheme, even if thelevel of the sidebands is higher. Furthermore, the total bandwidth ofthe radio frequency signal for the class B amplifier with pre-distortionmight well need to be larger than the combined radio frequency and biasbandwidths of the new system for achieving linearity. This is important,since not only does the new system split the bandwidth requirement intwo parts, which by itself makes it easier to find D/A converters thatfit, but it also lessens the total requirement for D/A bandwidth.

In the case of an input signal is in digital, complex number form, thesquared amplitude is easily calculated as the sum of the squaredin-phase (I) and quadrature-phase (Q) components. FIG. 9 illustrates adetail of a possible system using the squared amplitude as the input toa stored-table lookup system for providing radio frequency drive andbias modifications. Preferably, the lookup system is improved byinterpolation. The components of the digital complex input signal aresummed quadratically in a squaring and summing unit 43 of the inputdetector 40. This squared amplitude is provided as a control signal 47as in input to a complex table 27 of the pre-distortion unit 20. Theoutput of the complex table 27 may be interpolations between tablevalues and provided to a complex multiplier 24, where the complex table27 output is multiplied with the input signal 35. The result is providedto a complex-to-real upconverter 29 supplied by a digital oscillator 28.The result is provided as the drive signal 26.

Similarly, the control signal 47 is also provided as an input to a realtable 37 of the bias signal generator. The output of the real table 37may be interpolations between table values and provided as the biassignal 36.

This arrangement is also easily integrated with an adaptation means 44.The tables are in such an arrangement preferably adapted to maintainlinearity and output power also when circuit parameters change due totemperature and drift. This adaption is performed by signalsrepresenting coefficient modifications of the coefficients stored in thelook-up tables. These coefficient modifications are provided from theadaptation means 44 to the complex table 27 and the real table 37,respectively.

FIG. 12 illustrates the main steps of an embodiment of a methodaccording to the present invention. The procedure starts in step 200. Instep 202, the instantaneous amplitude of the input signal, or a quantityrelated thereto and derivable therefrom, is detected. A drive signal isderived from the input signal in step 204. This drive signal deriving ispreferably a pre-distortion of the input signal, dependent on thedetermined amplitude. In step 206, a bias signal is generated, whichalso is dependent on the determined amplitude. In step 208, the drivesignal is amplified using the bias signal level to provide an outputsignal. The procedure ends in step 214.

Thus, by using the properties of the projected amplitude probabilitydistribution, it is possible to perform modifications in drive and biassignals that reduces the radio frequency drive bandwidth expansion andincreases output power relative to the pre-distorted class B amplifier,and increases efficiency and output power compared to constant-gaindynamic biasing schemes.

When quasi-linear transconductance devices are used, the same ideas canbe used as for the system of FIG. 3. Usually, the maximum power isobtained by peaking in class A, but if the constant part of thetransconductance curve is large, peaking in class AB bias might give thehighest output power. This is also the case for trulyconstant-transconductance devices. The method according to the presentinvention can easily be modified to provide peaking in the highestoutput power class AB mode.

By allowing both the pre-distortion and the bias signal to vary withinput signal amplitude, one additional degree of freedom is achieved,which can be used in order to optimize the system behavior in differentaspects. When bandwidth is a limiting factor, it may be wise to firstselect a suitable pre-distortion function, having the desired bandwidth.Thereafter, the bias signal can be adjusted on order to provide therequired output characteristics. However, the opposite procedure is alsopossible, i.e. first determine a suitable bias signal function and thenadapt the pre-distorted drive signal according to bias signal. It isfurthermore possible to adapt both the bias signal function and thepre-distorted drive signal to achieve the required properties.

In the previous examples of the new system, some specific choices havebeen made. For example, the bandwidth of the drive signal has beenforced to stay within three times the input signal bandwidth. In casewider bandwidth can be afforded in the D/A converters and radiofrequency chains this can be used to improve efficiency. Higher averageefficiency can be obtained if the class AB and A regions for low andhigh amplitudes can be made smaller, thus increasing the class C region.This requires the use of a more wideband radio frequency drive signal.

The limitation of bandwidth expansion for the radio frequency drivesignal does not have to be carried out by explicitly using only low andodd powers of the input amplitude. In practice, higher powers, andfunctions not described by a power series, can be allowed, provided thatthe signal has little energy outside the allowed frequency range.

If one of the improvements that are possible with the new system is moreimportant than the others, only a part of the ideas may be needed. Thiscan in some cases reduce the complexity, or give improved performance.For example, using the highest output power class A (or AB) biasing inthe high end can be combined with class B, root function pre-distortionin the low end, with or without using an intermediate class C region. Inthis case the ideas are to be applied to an already existing system. Thestrategy is to make the fullest use of the bandwidth capabilities in thesystem, to improve efficiency, output power, or both.

An issue related to maximum power is the output power when the amplifieris overdriven. Overdrive is reflected in the output current waveform asa flattening of, or sometimes even a dip in, its uppermost part. Thiswaveform can contain more fundamental radio frequency current than thenon-flattened waveform, and so can give some increase in output power.To make use of this potential output power increase, drive and biaslevel modifications that together give linear output in the overdrivenregion must be determined. To limit the bandwidth expansion for thiscase, it may be useful to use another bias signal shape in the high endof the non-overdriven amplitude region, to go more smoothly into theoverdriven region. It should also be noted that more harmonics aregenerated under overdriven conditions, which can be detrimental to thesystem.

The ideas used in the present invention can also be used for generatinge.g. the non-linear currents of a Doherty “auxiliary” amplifier, forimproving such a system to cope with linear-transconductance devices andincreasing output power. A typical embodiment of a Doherty amplifierarrangement 60 is illustrated in FIG. 10. A radio frequency signal issimultaneously provided to a main amplifier 62 and an auxiliaryamplifier 64. The drive signal to the auxiliary amplifier 64 is modifiedin a controller 66 in order to achieve an operation of the auxiliaryamplifier 64 (also called peaking amplifier) only at high amplitudes.The output signals are combined with a quarter wavelength difference 68to provide the output signal to the transmitter 4. The particulardetails of the arrangement is not of particular importance for thepresent invention, but the important feature is the operation of theauxiliary amplifier 64. The operation is performed as a function of theinput signal amplitude, whereby the auxiliary amplifier 64 will start togive a zero output, and only above a certain amplitude give a linearoutput.

Here, the most important thing is to reduce the average DC currentconsumption, while at the same time obtain the maximum output radiofrequency current for the signal peaks. This means that, for all linearand most quasi-linear transconductance transistors, class A should beused at the peaks. Since ideally no radio frequency current should beproduced at amplitude levels below a certain point in amplitudedimension, the device can be biased in class C at low amplitude levels.I prior art, class C is typically used all the time for linear andquasi-linear transconductance transistors which limits output power andefficiency.

In FIG. 11, drive and bias signals of an embodiment an auxiliaryamplifier of a Doherty amplifier arrangement according to the presentinvention are illustrated. It can thus be noted that even by keeping thedrive signal essentially linear, a bias signal according to the figurecan give the requested output signal behavior. The low-order drivesignal results in a bandwidth essentially equal to the input signalbandwidth. In an amplitude range 14, the output signal is essentiallyzero. This is achieved by reducing the bias level considerably. The biassignal is kept below the amplitude-averaged bias signal level 10 in theamplitude range 12, and is allowed to increase in the amplitude range 11to give an almost linear rise of the output signal for largeramplitudes.

The signals according to FIG. 11 can for instance deliver 18% more radiofrequency current during the peaks, compared to a typical pre-distortedclass B arrangement, due to the use of class A biasing in this end.Furthermore, a pre-distorted class B arrangement has a very abrupt onsetof the drive signal, which causes a wide spectral bandwidth.

It is thus immediately realized that the principles according to thepresent invention can be used on a wide variety of systems. Differentsingle amplifier elements having different characteristics can be used,and the amplifiers can also be used in more elaborate amplifierarrangements. Non-exclusive examples are Doherty amplifier arrangements,Chireix amplifier arrangements and amplifier arrangements using envelopeelimination and restoration techniques.

Other non-linear current functions can also be generated with theproposed methods. Generally, maximum output power and efficiency can beimproved compared to the other systems, but in case large phasevariations with amplitude are to be incorporated, large reductions indrive signal bandwidth might not be feasible.

The dynamic bias used in the present invention gives generally anovercompensation of gain, i.e. the gain generally increases for higheramplitudes. In the embodiments described above, pre-distortion has beenused to compensate this gain variation caused by the dynamic bias. Asimilar result can also be achieved by instead using feedbackarrangements to modify the input signal into a drive signal. Thefeedback arrangements as such are known in prior art and are not furtherdescribed.

The invention disclosed herein makes it possible to get the maximumoutput power, or output current, from a practical radio frequencytransistor, with high efficiency and good linearity. The system improvesoutput power and efficiency compared to a constant-gain dynamic biasingsystem, and improves useful bandwidth and output power compared to apre-distorted class B amplifier. The improvements are fairlyinexpensive. Increased bandwidth for the bias signal is the maincomplication, and for this a higher efficiency, more output power andreduced and partitioned D/A-converter bandwidth are achieved. The sameideas that are used for making efficient linear amplifiers can also beused for the nonlinear “peaking” amplifier in a Doherty amplifiersystem, as well as for creating other more or less non-linear currentfunctions required in amplifier systems.

The system can be made more robust than prior art systems, since thepossibility of achieving significantly more output power with retainedor increased efficiency facilitates the inclusion of margins for circuiterrors. The reduced generation of harmonics in the high-power end alsolessens the requirements for repeatable circuit characteristics at theharmonic frequencies, which increases the production yield.

It will be understood by those skilled in the art that variousmodifications and changes may be made to the present invention withoutdeparture from the scope thereof, which is defined by the appendedclaims.

REFERENCES

[1] “RF Power Amplifiers for Wireless Communications” by S. C. Cripps,Artech House, Boston, 1999, pp. 45-60.

[2] “Output Performance of Idealized Microwave Power Amplifiers” by L.J. Kushner, Microwave Journal, Oct. 1989, pp. 103-116.

[3] “Low dissipation power and high linearity PCS power amplifier withadaptive gate bias control circuit” by K.-J. Youn et al., ElectronicsLetters, Vol. 32, No. 17, 15 Aug. 1996, pp. 1533-1535.

[4] “42% High-Efficiency Two-Stage HBT Power-Amplifier MMIC for W-CDMACellular Phone Systems” by T. Iwai et. al., IEEE Trans. MTT, Vol. 48,No. 12, December 2000, pp. 2567-2572.

[5] U.S. Pat. No. 5,923,215.

[6] U.S. Pat. No. 6,028,477.

[7] U.S. Pat. No. 5,808,511.

[8] “Improving the Power-Added Efficiency of FET Amplifiers Operatingwith Varying-Envelope Signals”, IEEE Trans. MTT, Vol. 31, No. 1, January1983, pp. 51-56.

[9] “A Thorough Investigation of Dynamic Bias on Linear GaAs FET PowerAmplifier Performance” by T. H. Miers, V. A. Hirsch, 1992 IEEE MTT-SDigest, pp. 537-540.

[10] “Increased Efficiency in QAM Power Amplifiers” by D. R. Conn, R. H.Hemmers, 1998 IEEE MTT-S Digest, pp. 1647-1650.

[11] “A Dynamic Efficient Bias Scheme Improves SSPA in AeronauticalSatellite Communication Systems” by I. K. Stubbs, IEEE Colloquium on‘Evolving Technologies for Small Earth Station Hardware’, Digest No.1995/037, IEEE, London, UK, 44 pp. p 5/1-8.

[12] “A New Adaptive Double Envelope Feedback (ADEF) Linearizer forSolid State Power Amplifiers”, IEEE Trans. MTT, Vol. 43, No. 7, July1995, pp. 1508-1515.

[13] U.S. Pat. No. 5,757,229.

[14] U.S. Pat. No. 6,130,579.

[15] European patent application 1 075 081.

1. Method of providing a radio frequency output signal, comprising the steps of: determining an instantaneous size measure of an input signal, said size measure being an amplitude or therefrom derivable quantity; deriving a drive signal from said input signal; providing a bias signal, being dependent on said instantaneous size measure; and amplifying said drive signal using a bias level according to said bias signal into said radio frequency output signal; whereby said bias signal dependency on said instantaneous size measure gives rise to an increased nonlinearity in said amplifying step.
 2. Method according to claim 1, whereby said bias signal gives an amplification according to one of class C and class B for instantaneous size measures within a first amplitude range, and said bias signal being higher than class B amplification for instantaneous size measures above said first amplitude range.
 3. Method according to claim 2, whereby said bias signal is controlled to give essentially a class A bias level at maximum amplitude.
 4. Method according to claim 1, whereby said bias signal providing step is controlled for producing a predetermined output characteristics, whereby a bias signal amplitude-averaged over an amplitude interval comprising all amplitudes in an entire amplitude range supported by said amplifying step above a first amplitude is higher than a bias signal amplitude-averaged over said entire amplitude range.
 5. Method according to claim 1, wherein said deriving step comprises the step of modifying said input signal.
 6. Method according to claim 5, wherein said deriving step comprises the step of pre-distorting said input signal dependent on said instantaneous size measure.
 7. Method according to claim 5, wherein said deriving step comprises the step of modifying said input signal by a feedback arrangement.
 8. Method according to claim 1, wherein said bias signal is controlled to, for all amplitudes within a first amplitude range, increase with increasing amplitude.
 9. Method according to claim 1, wherein said bias signal is controlled to be, for all amplitudes within a second amplitude range, lower than said bias signal amplitude-averaged over said entire amplitude range.
 10. Method according to claim 8, wherein said first amplitude range comprises maximum amplitude.
 11. Method according to claim 6, comprising the further steps of: selecting a pre-distortion function having a predetermined bandwidth; and adapting bias signal according to said pre-distortion function.
 12. Method according to claim 11, wherein said pre-distortion function contains predominantly low-order components.
 13. Method according to claim 6, comprising the further steps of: selecting said bias signal according to predetermined relations; and adapting said pre-distortion function according to said bias signal.
 14. Method according to claim 1, wherein said output characteristics, at least for a third amplitude range, is linear.
 15. Method according to claim 14, wherein said output characteristics is substantially linear over the entire amplitude range.
 16. Method according to claim 1, wherein said output characteristics comprises a substantially zero output signal within a fourth amplitude range.
 17. Method according to claim 1, comprising the further steps of: determining a feedback signal of said radio frequency output signal; and adapting said drive signal and/or said bias signal according to said feedback signal.
 18. Method according to claim 6, comprising the further steps of: causing said pre-distorting and bias signal providing steps to be simultaneous at the input of said amplification.
 19. Method according to claim 18, wherein said causing step in turn comprises at least one of the steps of: inverse filtering of said drive signal with respect to a first signal path to an amplifying element; delay compensation of said drive signal with respect to said first signal path to an amplifying element; inverse filtering of said bias signal with respect to a second signal path to said amplifying element; and delay compensation of said bias signal with respect to said second signal path to said amplifying element.
 20. Method according to claim 1, comprising the further step of: compensating current saturation at high amplitude end.
 21. Use of a method according to claim 1 in a radio frequency amplifier arrangement of a type selected from the list of: Doherty amplifier arrangement; Chireix amplifier arrangement; and amplifier arrangements using envelope and restoration enhancement techniques.
 22. Radio frequency power amplifier, comprising: input signal terminal; input detector arranged to determine an instantaneous size measure of a signal on said input signal terminal, said size measure being an amplitude or therefrom derivable quantity; drive signal deriving means connected to said input signal terminal, providing a drive signal; bias signal generator providing a bias signal, said bias signal generator being connected to said input detector and being controlled dependent on said instantaneous size measure; and amplifying element, connected to said drive signal deriving means and said bias signal generator; whereby said bias signal generator being controlled to gives rise to an increased nonlinearity in said amplifying element.
 23. Radio frequency power amplifier according to claim 22, wherein said bias signal generator is arranged to give an amplification in said amplifying element according to one of class C and class B for instantaneous size measures within a first amplitude range, and to give a bias signal being higher than class B amplification for instantaneous size measures above said first amplitude range.
 24. Radio frequency power amplifier according to claim 22, wherein said bias signal generator is arranged to give a bias signal amplitude-averaged over an amplitude interval comprising all amplitudes in an entire amplitude range supported by said amplifying element above a first amplitude is higher than a bias signal amplitude-averaged over said entire amplitude range.
 25. Radio frequency power amplifier according to claim 22, wherein said drive signal deriving means comprises pre-distorting means connected to said input detector, being controlled dependent on said instantaneous size measure.
 26. Radio frequency power amplifier according to claim 22, wherein said bias signal generator in turn comprises means giving a bias signal, which for all amplitudes within a first amplitude range, increase with increasing amplitude.
 27. Radio frequency power amplifier according to claim 22, wherein said bias signal generator in turn comprises means giving a bias signal, which for all amplitudes within a second amplitude range, is lower than an amplitude-averaged bias signal.
 28. Radio frequency power amplifier according to claim 25, further comprising: feed-back arrangement, in turn comprising a feedback sensor monitoring said output of said amplifier element and adaptation means connected said bias signal generator and said pre-distortion means for providing said bias signal generator and said pre-distortion means with a feedback signal; said bias signal generator and said pre-distortion means being arranged to adapt their actions according to said feedback signal.
 29. Radio frequency power amplifier according to claim 22, further comprising: simultaneousness-causing means for causing said drive signal and bias signal to be simultaneous at in input of said amplifying element.
 30. Radio frequency power amplifier according to claim 29, wherein said coincidence causing means in turn comprises at least one of: inverse filter connected between said pre-distortion means and said amplifying element, for compensating for a first signal path to said amplifying element; and inverse filter connected between said bias signal generator and said amplifying element, for compensating for a second signal path to said amplifying element.
 31. Composite radio frequency power amplifier, comprising at least one radio frequency power amplifier according to claim 22 as a sub-amplifier.
 32. Composite radio frequency power amplifier according to claim 31, wherein said composite radio frequency power amplifier is selected from the list of: Doherty amplifier arrangement; Chireix amplifier arrangement; and amplifier arrangements using envelope elimination and restoration techniques.
 33. Transmitter, having a radio frequency power amplifier, said radio frequency power amplifier comprising: input signal terminal; input detector arranged to determine an instantaneous size measure of a signal on said input signal terminal, said size measure being an amplitude or therefrom derivable quantity; drive signal deriving means connected to said input signal terminal, providing a drive signal; bias signal generator providing a bias signal, said bias signal generator being connected to said input detector and being controlled dependent on said instantaneous size measure; and amplifying element, connected to said drive signal deriving means and said bias signal generator; whereby said bias signal generator being controlled to gives rise to an increased nonlinearity in said amplifying element.
 34. Transmitter according to claim 33, wherein said bias signal generator is arranged to give an amplification in said amplifying element according to one of class C and class B for instantaneous size measures within a first amplitude range, and to give a bias signal being higher than class B amplification for instantaneous size measures above said first amplitude range.
 35. Transmitter according to claim 33, wherein said bias signal amplitude-averaged over an amplitude interval comprising all amplitudes in an entire amplitude range supported by said amplifying element above a first amplitude is higher than a bias signal amplitude-averaged over said entire amplitude range.
 36. Transmitter according to claim 33, wherein said drive signal deriving means comprises pre-distorting means connected to said input detector, being controlled dependent on said instantaneous size measure.
 37. Transmitter according to claim 33, wherein said bias signal generator in turn comprises means giving a bias signal, which for all amplitudes within a first amplitude range, increase with increasing amplitude.
 38. Transmitter according to claim 33, wherein said bias signal generator in turn comprises means giving a bias signal, which for all amplitudes within a second amplitude range, is lower than an amplitude-averaged bias signal.
 39. Transmitter according to claim 38, wherein said second amplitude range covers at least half the amplitude distribution.
 40. Transmitter according to claim 38, wherein said pre-distortion means comprises means for making said drive signal larger than said input signal at least in said second amplitude range.
 41. Wireless communication system, having a radio frequency power amplifier, said radio frequency power amplifier comprising: input signal terminal; input detector arranged to determine an instantaneous size measure of a signal on said input signal terminal, said size measure being an amplitude or therefrom derivable quantity; drive signal deriving means connected to said input signal terminal, providing a drive signal; bias signal generator providing a bias signal, said bias signal generator being connected to said input detector and being controlled dependent on said instantaneous size measure; and amplifying element, connected to said drive signal deriving means and said bias signal generator; whereby said bias signal generator being controlled gives rise to an increased nonlinearity in said amplifying element.
 42. Base station of a wireless communication system, having a radio frequency power amplifier, said radio frequency power amplifier comprising: input signal terminal; input detector arranged to determine an instantaneous size measure of a signal on said input signal terminal, said size measure being an amplitude or therefrom derivable quantity; drive signal deriving means connected to said input signal terminal, providing a drive signal; bias signal generator providing a bias signal, said bias signal generator being connected to said input detector and being controlled dependent on said instantaneous size measure; and amplifying element, connected to said drive signal deriving means and said bias signal generator; whereby said bias signal generator being controlled gives rise to an increased nonlinearity in said amplifying element.
 43. Mobile unit of a wireless communication system, having a radio frequency power amplifier, said radio frequency power amplifier comprising: input signal terminal; input detector arranged to determine an instantaneous size measure of a signal on said input signal terminal, said size measure being an amplitude or therefrom derivable quantity; drive signal deriving means connected to said input signal terminal, providing a drive signal; bias signal generator providing a bias signal, said bias signal generator being connected to said input detector and being controlled dependent on said instantaneous size measure; and amplifying element, connected to said drive signal deriving means and said bias signal generator; whereby said bias signal generator being controlled gives rise to an increased nonlinearity in said amplifying element. 